Software defined automotive radar

ABSTRACT

A method for operating a radar sensing system includes configuring a transmitter to transmit a radio signal. A receiver is configured to receive radio signals. The received radio signals include the transmitted radio signal transmitted by the transmitter and reflected from objects in the environment. The method includes with advanced temporal knowledge of the codes used to modulate the transmitted radio signal, using code values of the plurality of codes, and in combination with a bank of digital finite impulse response (FIR) filters, generating complementary signals of any self-interference noise. The method further includes subtracting the complementary signals at one or more points in the receiver prior to the interference desensing the receiver. The radar sensing system further includes a frequency modulated continuous wave (FMCW) interference canceller for detecting the largest interference signals and sequentially cancelling them while signal processing the received radio signals.

CROSS REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. application Ser. No.17/683,711, filed Mar. 1, 2022, now U.S. Pat. No. 11,614,538, which is acontinuation of U.S. patent application Ser. No. 16/383,950, filed Apr.15, 2019, now U.S. Pat. No. 11,262,448, which is a continuation of U.S.patent application Ser. No. 15/496,038, filed Apr. 25, 2017, now U.S.Pat. No. 10,261,179, which claims the filing benefits of U.S.provisional applications, Ser. No. 62/327,003, filed Apr. 25, 2016, Ser.No. 62/327,004, filed Apr. 25, 2016, Ser. No. 62/327,005, filed Apr. 25,2016, Ser. No. 62/327,006, filed Apr. 25, 2016, Ser. No. 61/327,015,filed Apr. 25, 2016, Ser. No. 62/327,016, filed Apr. 25, 2016, Ser. No.62/327,017, filed Apr. 25, 2016, and Ser. No. 62/327,018, filed Apr. 25,2016; and U.S. Pat. No. 10,261,179 is a continuation-in-part of U.S.patent application Ser. No. 15/481,648, filed Apr. 7, 2017, now U.S.Pat. No. 9,689,967, which claims the filing benefits of U.S. provisionalapplications, Ser. No. 62/319,613, filed Apr. 7, 2016, and Ser. No.62/327,003, filed Apr. 25, 2016, which are all hereby incorporated byreference herein in their entireties.

FIELD OF THE INVENTION

The present invention is directed to radar systems, and in particular toradar systems for vehicles.

BACKGROUND OF THE INVENTION

The use of radar to determine range, velocity, and angle (elevation orazimuth) of objects in an environment is important in a number ofapplications including automotive radar and gesture detection. A radartypically transmits a radio frequency (RF) signal and listens for thereflection of the radio signal from objects in the environment. A radarsystem estimates the location and velocity of objects, also calledtargets, in the environment by comparing the received radio signal withthe transmitted radio signal. It would be advantageous to have a radarsystem that can adapt various aspects of the radar transmitted signaland receiver processing to different environments and differentobjective functions.

SUMMARY OF THE INVENTION

The present invention provides methods and a radar system that canoperate under a variety of environments, a variety of externalinformation, and with a variety of objective functions to modify thetransmission and reception processing at a given time to optimize thesystem with respect to a given objective function. The inventionaccomplishes better performance by adaptively changing the systemincluding changing the transmitted signal characteristics such as thebaseband signal, the bandwidth, the frequency, and the power and the setof transmitting antennas that are used. Better performance is alsoobtained by changing the receiver processing including the receivingantennas, interference mitigation techniques to be employed, length oftime of the signal used to process a received signal to determine range.

A radar sensing system for a vehicle in accordance with an embodiment ofthe present invention includes at least one transmitter, at least onereceiver, at least one antenna, memory, and a control processor. The atleast one transmitter is configured for installation and use on avehicle and is operable to or configured to transmit a radio signal. Theat least one transmitter is further operable to transmit radio signals.The transmitted radio signals are generated by up-converting a basebandtransmitted signal. The at least one receiver is configured forinstallation and use on the vehicle and is operable to or configured toreceive reflected radio signals. The reflected radio signals are thetransmitted radio signals reflected from an object or multiple objects.The radar system includes one or more receivers. In each receiver thereceived reflected radio signal is down-converted (with in-phase andquadrature signals), and then sampled and quantized using ananalog-to-digital converter (ADC) to produce possibly complex basebandsamples. The resulting complex signal from the ADC is processed by adigital processor. A control unit is employed to change thecharacteristics of the transmitted signal and in the way the receiverprocesses the reflected RF signal to generate estimates of range,velocity, and angle of objects in the environment.

A radar sensing system for a vehicle in accordance with an embodiment ofthe present invention, the radar sensing system includes a plurality oftransmitters, a plurality of receivers, and a control unit. Theplurality of transmitters are configured for installation and use on avehicle, and operable to or configured to transmit modulated radiosignals. The plurality of receivers are configured for installation anduse on the vehicle, and operable to or configured to receive radiosignals. The received radio signals are transmitted radio signalsreflected from an object in the environment. Each transmitter of theplurality of transmitters comprises a digital processing unit, adigital-to-analog converter, an analog processing unit, and transmittingantennas. Each receiver of the plurality of receivers comprises areceiving antenna, an analog processing unit, an analog-to-digitalconverter, and a digital processing unit. The control unit is furtheroperable to or configured to individually modify one or moretransmitters of the plurality of transmitters and one or more receiversof the plurality of receivers. The control unit may be operable to orconfigured to select an operating mode from a single-input,multiple-output (SIMO) mode and a multi-input, multiple-output (MIMO)mode.

These and other objects, advantages, purposes and features of thepresent invention will become apparent upon review of the followingspecification in conjunction with the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a plan view of an automobile equipped with a radar system inaccordance with the present invention;

FIGS. 2A and 2B are block diagrams of single transmitter and receiver ina radar system;

FIG. 3 is a block diagram of multiple transmitters and multiplereceivers in a radar system;

FIG. 4 is a block of a single receiver and single transmitter;

FIG. 5 is a graph illustrating an exemplary transmitted signal using anm-sequence of length 31 in accordance with the present invention;

FIGS. 6-9 are graphs illustrating exemplary matched filter outputs overtime in accordance with the present invention;

FIG. 10 is a graph illustrating an exemplary imagery part of filteroutput vs a real part of filter output in accordance with the presentinvention;

FIGS. 11 a, 11 b, and 11 c are block diagrams illustrating exemplarysteps to signal processing in accordance with the present invention;

FIG. 12 is a block diagram of an exemplary controller interacting with areceiver and transmitter of a radar system in accordance with thepresent invention; and

FIGS. 13A and 13B are block diagrams of an exemplary radar systemarchitecture with multiple receivers in accordance with the presentinvention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention will now be described with reference to theaccompanying figures, wherein numbered elements in the following writtendescription correspond to like-numbered elements in the figures. Methodsand systems of the present invention may achieve better performance froma radar system when there is a near object and a far object. Exemplaryembodiments of the present invention accomplish better performance byadjusting the radar system to the environment, the objective and inputsexternal to the radar system. The invention accomplishes betterperformance by adapting the radar system under software control.

The radar sensing system of the present invention may utilize aspects ofthe radar systems described in U.S. Pat. Nos. 9,575,160 and/or9,599,702, and/or U.S. patent application Ser. No. 15/416,219, filedJan. 26, 2017, now U.S. Pat. No. 9,772,397, and/or Ser. No. 15/292,755,filed Oct. 13, 2016, now U.S. Pat. No. 9,753,121, and/or U.S.provisional applications, Ser. No. 62/382,857, filed Sep. 2, 2016,and/or Ser. No. 62/381,808, filed Aug. 31, 2016, which are all herebyincorporated by reference herein in their entireties.

As illustrated in FIG. 1 , there may be multiple radars (e.g., 104 a-104d) embedded into an automobile. Each of these could employ the ideascontained in the present invention. FIG. 1 illustrates an exemplaryradar system 100 configured for use in a vehicle 150. In an aspect ofthe present invention, a vehicle 150 may be an automobile, truck, orbus, etc. As illustrated in FIG. 1 , the radar system 100 may compriseone or more transmitters and one or more virtual receivers 104 a-104 d,control and processing module 102 and indicator 106. Otherconfigurations are also possible. FIG. 1 illustratesreceivers/transmitters 104 a-104 d placed to acquire and provide datafor object detection and adaptive cruise control. The radar system 100(providing such object detection and adaptive cruise control or thelike) may be part of an Advanced Driver Assistance System (ADAS) for theautomobile 150.

A radar system operates by transmitting a signal and then listening forthe reflection of that signal from objects in the environment. Bycomparing the transmitted signal and the received signal, estimates ofthe range to different objects, the velocity of different objects andthe angle (azimuth and/or elevation) can be estimated.

There are several different types of signals that transmitters in radarsystems employ. A radar system may transmit a continuous signal or apulsed signal. In a pulsed radar system the signal is transmitted for ashort time and then no signal is transmitted. This is repeated over andover. When the signal is not being transmitted the receiver listens forechoes or reflections from objects in the environment. Often a singleantenna is used for both the transmitter and receiver and the radartransmits on the antenna and then listens to the received signal on thesame antenna. This process is then repeated. In a continuous wave radarsystem the signal is continuously transmitted. There may be an antennafor transmitting and a separate antenna for receiving. One type ofcontinuous wave radar signal is known as frequency modulated continuouswave (FMCW) radar signal. In FMCW the transmitted signal is a sinusoidalsignal with varying frequency. By measuring the time difference betweenwhen a certain frequency was transmitted and when the received signalcontained that frequency the range to an object can be determined.

A second type of continuous wave signal used in radar systems is a phasemodulated continuous wave (PMCW) signal. In this type of radar system,the transmitted signal is a sinusoidal signal in which the phase of thesinusoidal signal varies. Typically, the phase during a given timeperiod (called a chip period or chip duration) is one of a finite numberof possible phases. A spreading code consisting of sequence of chips,(e.g., +1, +1, −1, +1, −1, . . . ) that is mapped (e.g., +1→0, −1→π)into a sequence of phases (e.g., 0, 0, π, 0, π, . . . ) that is used tomodulate a carrier to generate the radio frequency (RF) signal. Thespreading code could be a periodic sequence or could be a pseudo-randomsequence with a very large period so it appears to be a nearly randomsequence. The spreading code could be a binary code (e.g., +1 or −1).The resulting signal has a bandwidth that is proportional to the rate atwhich the phases change, called the chip rate, which is the inverse ofthe chip duration. By comparing the return signal to the transmittedsignal the receiver can determine the range and the velocity ofreflected objects.

There are several ways to implement a radar system. One way, shown inFIG. 2A uses a single antenna 202 for transmitting and receiving. Theantenna is connected to a duplexer 204 that routes the appropriatesignal from the antenna to the receiver (208) or routes the signal fromthe transmitter 206 to the antenna 202. A control processor 210 controlsthe operation of the transmitter and receiver and estimates the rangeand velocity of objects in the environment. A second way to implement aradar system is shown in FIG. 2B. In this system there are separateantennas for transmitting (202A) and receiving (202B). A controlprocessor 210 performs the same basic functions as in FIG. 2A. In eachcase there may be a display to visualize the location of objects in theenvironment.

A radar system with multiple antennas, transmitters and receivers isshown in FIG. 3 . Using multiple antennas allows a radar system todetermine the angle (azimuth or elevation or both) of targets in theenvironment. Depending on the geometry of the antenna system differentangles (e.g., azimuth or elevation) can be determined.

The radar system may be connected to a network via an Ethernetconnection or other types of network connections 314. The radar systemwill have memory (310, 312) to store software used for processing thesignals in order to determine range, velocity and location of objects.Memory can also be used to store information about targets in theenvironment.

A basic block diagram of a PMCW system with a single transmitter andreceiver is shown in FIG. 4 . The transmitter 400, as shown in FIG. 4 ,consists of a digital signal generator 410, followed by adigital-to-analog converter (DAC) 420. The output of the DAC followed isup converted to a RF signal and amplified by the analog processing 430unit. The result is then used as the antenna 440 input. The digitalsignal generator generates a baseband signal. The receiver, as shown inFIG. 4 , consists of a receiving antenna 460, an analog processing unitthat down amplifies the signal and mixes the signal to baseband 470.This is followed by an analog-to-digital converter (ADC) 480 and thendigital baseband processing 490. There is also a control processor (notshown) that controls the operation of the transmitter and receiver. Thebaseband processing will process the received signal and may generatedata that can be used to determine range, velocity and angle.

Radars must operate in various environments. For example, an automotiveradar must operate in urban areas, suburban areas, rural areas, rain,snow, deserts, parking lots, garages, construction zones, to name a few.Depending on the installation location of the radar in an automobile,the transmitted signal might be reflected off of parts of theautomobile. For example, reflections from a bumper in the automobilemight create very strong self-interference. The set of environments anautomobile is expected to operate in is extensive. Depending on theenvironment different types of signals might be used. A radar signalappropriate for one environment will not be the best signal to use in adifferent environment. The receiver processing used will also depend onthe environment. The environment might be determined from the radaritself but also could be obtained by the radar from external sources(e.g., other vehicles, cellular networks, GPS).

In addition to operating in multiple environments, radar systems mayhave different performance objectives. Range resolution, maximumunambiguous range, Doppler resolution, angular resolution, and field ofview are some of the objectives of a radar system. The smallestseparation of two objects, such that they are recognized as two distinctobjects by a radar, is known as the range resolution of the radar. Therange resolution is inversely proportional to the bandwidth of thetransmitted signal. A short-range radar (SRR) might provide a rangeresolution that is sub-meter (e.g., less than 5 cm) but only fordistances from 0 to less than 30 meters. A long-range radar might have amuch larger range resolution. Another performance measure is the maximumunambiguous range, D_(u). This is the maximum distance of an object suchthat the distance can be correctly (unambiguously) determined from thereceived (reflected) signal. If the delay of the reflected signal can beconfused with another (shorter) delay due to the period of thetransmitted signal, then the distance to the object cannot beunambiguously determined. A long-range radar (LRR) might have a maximumunambiguous range out to several hundred meters whereas a SRR might havean unambiguous range out to several tens of meters.

Doppler resolution refers to the capability of a radar to discriminatethe velocity of different targets. There is a maximum Doppler shift thata radar can determine without ambiguity. This is known as the maximumunambiguous velocity. A radar system using multiple antennas candetermine the angle of a target relative to some reference in either thehorizontal plane (azimuth) or the elevation angle (angle relative to thehorizontal plane). A set of angles for which a radar can detect anobject is called the field of view. Generally, with a fixed number ofantennas, a large field of view would result is less angular resolutionwhile a narrow field of view can provide better angular resolution. Withcertain antenna configurations, the elevation angle of an object can bedetermined.

The description herein includes a radar system in which there are N_(T)transmitters and NR receivers N_(T)×N_(R) virtual radars, one for eachtransmitter-receiver pair. For example, a radar system with eighttransmitters and eight receivers will have 64 pairs or 64 virtual radars(with 64 virtual receivers). When three transmitters (Tx1, Tx2, Tx3)generate signals that are being received by three receivers (Rx1, Rx2,Rx3), each of the receivers is receiving the transmission from each ofthe transmitters reflected by objects in the environment. Each of thereceivers is receiving the sum of reflected signals due to all three ofthe transmissions at the same time. Each receiver can attempt todetermine the range and Doppler of objects by correlating with delayedreplicas of the signal from one of the transmitters. The physicalreceivers may then be “divided” into three separate virtual receivers,each virtual receiver correlating with a replica of one of thetransmitted signals. In a preferred radar system of the presentinvention, there are 1-4 transmitters and 4-8 receivers, or morepreferably 4-8 transmitters and 8-16 receivers, and most preferably 16or more transmitters and 16-64 or more receivers.

As mentioned earlier, there are various types of signals used in radarsystems. A pulsed radar transmits a signal for a short duration of timethen turns off the transmitter and listens for reflections. A continuouswave radar transmits a continuous signal. One type of continuous waveradar signal is known as frequency modulated continuous wave (FMCW)signal. The frequency of this signal is varied from some low frequencyvalue to a high frequency value over some time interval and thenrepeats. Another type of continuous wave radar signal is known as phasemodulated continuous wave (PMCW). The phase of the transmitted signal isvaried in PMCW. Often the variation of the phase is according to aspreading code. The spreading code may be binary (e.g., +1 and −1) inwhich case the phase of the transmitted signal at any time takes on oneof two possible values (e.g., 0 and π radians). Spreading codes withmore than two levels can also be used. Often the code repeats after acertain duration in time duration, sometimes called the pulse repetitioninterval (PRI). Various types of spreading codes can be used. Theseinclude pseudorandom binary sequence (PRBS) codes also calledm-sequences, almost perfect autocorrelation sequences (APAS), Golaycodes, constant amplitude zero autocorrelation codes (CAZAC) also knownas Frank-Zadoff-Chu (FZC) sequences, as well as many other codes thatcan be used. In a radar system with a single antenna, a single spreadingcode is used. The autocorrelation of this single code determines thecapability of the radar to estimate the range (range resolution andmaximum unambiguous range). Codes with good autocorrelation propertiesinclude Barker sequences, m-sequences, FZC sequences, and Golay codes.These codes have small sidelobes (the off-center autocorrelation). Codesthat have ideal autocorrelation (e.g., Golay codes, CAZAC) can haverange sidelobes in the presence of non-zero Doppler shift that willlimit the detectability of far targets in the presence of near targets.

In a multiple-input, multiple-output (MIMO) system, there are multipletransmitters that operate simultaneously. Each transmitter uses aspreading code and thus multiple codes are needed, one for eachtransmitter. In this case (multiple transmitters), codes that have goodautocorrelation, as well as good cross correlation properties aredesirable. Generally, the better the autocorrelation of codes, the worsethe cross correlation properties.

FIG. 5 shows a baseband signal which has a period of L_(C)=31. The chipsin this example are from a maximal length sequence (m-sequence) oflength L_(C)=31 generated by an exemplary shift register of length 5.Note that the signal repeats every L_(C) chips or L_(C)T_(C) seconds.The pulse repetition rate is R_(PR)=1/(L_(C)T_(C)). The transmittedsignal is generated from the baseband signal by modulating the basebandsignal onto a carrier frequency to generate a radio frequency signal.

As illustrated in FIG. 4 , the received signal is down-converted to acomplex baseband signal via an RF front end analog signal processing470. The analog signal processing involves amplification, mixing with alocal oscillator signal, and filtering. The mixing is with twosinusoidal signals that are 90 degrees out of phase (e.g., cosine andsine or in-phase and quadrature-phase signals). After down conversion,the complex analog baseband signal is converted to a complex basebanddigital signal by using analog-to-digital converters (ADCs) 480. Thecomplex baseband digital signal (output by the ADCs 480) is then theinput to a digital processing unit 490. The digital processing unit 490performs correlations or matched filtering. The correlators multiply thereceived complex baseband signal by a delayed replica of the basebandtransmitted signal and then the result is accumulated over a certaintime interval. A bank of correlators where each correlator has adifferent delay used for the replica of the baseband transmitted signalwill produce a set of correlations that correspond to different rangesof objects. In essence, a correlator that has a particular delay of thebaseband transmitted signal is looking for the presence of a reflectionfrom an object at a distance corresponding to the particular delay forthe particular correlator, and for which the round-trip delay is thedelay used for the baseband transmitted signal.

A matched filter is a device that produces all correlations for allpossible delays. That is, the output of the matched filter at a giventime corresponds to a correlation with a given delay applied to thetransmitted signal when doing the correlation. The matched filterprovides all possible correlations. Note that the matched filter shouldproduce a complex output because the input is complex. Alternatively,there could be a filter for the real part of the input and a filter forthe imaginary part of the input. A matched filter can also beimplemented by a fast Fourier transform (FFT) of the received complexbaseband signal and the corresponding transmitted signal, multiplyingthe results, and then taking an inverse fast Fourier transform (IFFT).

FIG. 5 illustrates a baseband signal which has a period of L_(C)=31. Thechips in this example are from a maximal length sequence (m-sequence) oflength L_(C)=31 generated by an exemplary shift register of length 5.Note that the signal repeats every L_(C) chips or L_(C)T_(C) seconds.The pulse repetition rate is R_(PR)=1/(L_(C)T_(C)). The transmittedsignal is generated from the baseband signal by modulating the basebandsignal onto a carrier frequency to generate a radio frequency signal.

FIG. 6 shows the real part of the output of a matched filter due to thetransmitted baseband signal shown in FIG. 5 . Here it is assumed theradar started to transmit at time 0 and there is no delay between thetransmitter and receiver. That is, there is an object at distance 0. Thematched filter output before a full period of the signal is transmittedgenerates partial correlations. That is, it correlates with only aportion of the code because only a portion of the code has beentransmitted. Only after the entire period of the code has beentransmitted does the correlation reach a peak. In continuous operation,an object that has a delay of one period of the spreading code willappear to have the same delay as an object at distance 0. Thus, a radarusing this system cannot determine whether the delay is 0, one period ofthe spreading code, two periods of the spreading code, and so on.Therefore, the maximum unambiguous range in this case corresponds to atmost one period of the spreading code. A longer spreading code willyield a larger maximum unambiguous range. A delay of τ corresponds to arange of τc/2 where c is the speed of light. The factor of two isbecause the delay corresponds to the round-trip time from the radar tothe target and back to the radar. Here the assumption is that thetransmitter and receiver are approximately co-located.

FIG. 7 illustrates the real part of the output of the matched filterwhen there are two objects that have a differential range delay of 2chip durations. The filter output shows two distinct peaks in the outputof the matched filter.

For PMCW radar systems that utilize nonideal spreading codes andcorrelate over a certain time interval, the autocorrelation is notideal. That is, the sidelobes are not zero. The sidelobes of a neartarget can mask the peak of the correlation for a far object or targetbecause the signal from the near object or target is far stronger thanthe signal from the far object or target.

Range Estimation

FIG. 8 illustrates the case where the differential round trip delaybetween two targets is one chip duration. In this case, two objectscannot be distinguished and thus the range resolution of this wouldcorrespond to the differential distance corresponding to a duration ofhalf (½) a chip. If Tc denotes the chip duration and Lc denotes thenumber of chips in one period of the transmitted sequence, then the rateat which the sequence is repeated is Rpr=1/(LcTc), which is sometimesreferred to as the pulse repetition rate even though this is acontinuous type of signal. If c denotes the speed of light, then therange resolution is given by:D _(R)=(T _(C)/2)c=c/(2R _(PR) L _(C)).If a signal repeats every T_(PR) or at rate R_(PR), then the maximumunambiguous range D_(U) is:D _(U) =cT _(PR)/2=(cT _(C) L _(C))/2=c/(2R _(PR)).Two targets separated by the maximum unambiguous range will appear tothe radar systems as being at the same range. This is sometimes calledrange aliasing. If the chip duration, T_(C), is decreased, then therange resolutions would improve proportionally. However, changing thechip duration changes the bandwidth, which might be limited byregulations. If there are 31 chips per period of the spreading code,there are at most 31 different ranges that can be distinguished. As anexample, if T_(C)=10 nanoseconds (a chiprate of 100 Mchips/second), thenthe range resolution would be limited to 1.5 meters. That is, twoobjects separated by less than 1.5 m would cause reflected signals to beless than a chip duration apart in delay. For this example, the maximumunambiguous range would be 46.5 m. That is, an object at a distance of46.5 m would cause a reflected signal to have a delay exactly equal tothe period of the signal and thus would appear as an object at adistance of 0 m. A longer spreading code would provide for a largerunambiguous range. For example, a spreading code of length 1023 wouldprovide a maximum unambiguous range of about 1,534 m.Velocity Estimation

Another goal of an exemplary radar system is to estimate thedifferential velocity between the radar system and a target. Becausetargets in the environment, or the radar itself, are moving, the signalreflected from an object will not have the same frequency as thetransmitted signal. This effect is known as the Doppler Effect and canbe used to determine the relative velocity of targets in theenvironment. Suppose the differential (radial) velocity of the targetrelative to the radar system is Δv and the carrier frequency is f_(C).Then, the Doppler frequency shift is f_(D)=2ΔV f_(C)/c. This is becausethere is a Doppler shift of ΔVf_(C)/c between the radar transmitter andthe target and then an additional ΔVf_(C)/c Doppler shift of thereflected signal from the target to the radar receiver. For example, acarrier frequency of 79 GHz with a differential velocity of 300km/hour=83.3 m/s would result in a frequency shift of about 44 kHz. Afrequency shift of f_(D) corresponds to a differential velocity ofΔV=(f_(D))c/(2f_(C)).

Suppose that a signal, for example an m-sequence, is repeated N times.This is called a scan. The period of the signal is L_(C)T_(C). The timeduration of the scan is N*L_(C)T_(C). During each repetition, acorrelation with a spreading code with a given delay (e.g.,corresponding to the delay with a given target) is calculated. Thiscorrelation calculation generates a complex number for a given delay andthis repeats N times during a scan. The N complex numbers can be used todetermine the Doppler frequency shift at the given delay. In the absenceof Doppler frequency shift the complex correlation values will beconstant. In the presence of a Doppler shift the complex correlationvalues will rotate. The rate of rotation will be related to the Dopplerfrequency. FIG. 9 illustrates the real and imaginary parts of thematched filter output when there is a Doppler shift. FIG. 10 illustratesthe complex values at the peak correlation outputs. As can be seen, thematched filter output is rotating around a circle. The rate of rotationis a measure of the Doppler frequency. Knowing the Doppler frequencyallows a calculation of the relative velocity of a target.

One way to estimate the Doppler frequency is to use a fast Fouriertransform (FFT) on the complex samples. With this approach to estimatingthe frequency shift due to Doppler, with N points as the input to theFFT, there will also be N frequency points generated. The frequencyresolution possible is over the range of frequencies from a negativefrequency of −R_(PR)/2 to a positive frequency+R_(PR)/2 or a range ofR_(PR). Thus, the spacing between frequency points will bef_(R)=R_(PR)/N. This is the frequency resolution. This corresponds to avelocity resolution of:V _(r) =cR _(pr)/(2f _(c) N).

If the complex correlation samples are produced at a rate ofR_(PR)=1/T_(PR)=1/L_(C)T_(C), then the frequency range that those pointsrepresent is limited to −R_(PR)/2 to +R_(PR)/2. Thus, the maximumunambiguous differential frequencies f_(u) that can be represented isgiven by −R_(pri)/2<f_(u)<+R_(pri)/2. When this is converted tovelocity, the result is that the maximum unambiguous velocity is limitedto values in the interval shown below:−cR _(PR)/(4f _(C))<V _(U) <+cR _(PR)/(4f _(C)).

Increasing the repetition rate increases the maximum unambiguousvelocities that can be determined. However, increasing the repetitionrate decreases the maximum unambiguous range that can be determined. Theproduct of the maximum unambiguous velocity and maximum unambiguousrange is limited as−c ²/(8f _(c))<D _(u) V _(u) <c ²/(8f _(c))which is independent of the various parameters of the transmittedsignal, except the carrier frequency.

The product of the velocity resolution and the range resolution is givenasD _(r) V _(r) =c{circumflex over ( )}2/(4_(FC) L _(C) N)where L_(C) is the number of chips in a single period of the spreadingcode and N is the number of points in the FFT used to determine thevelocity. For a fixed scan time (L_(C)N T_(C)) and fixed chip durationT_(C), there is a tradeoff between the resolution possible for the rangeand the resolution possible for the velocity. By increasing N anddecreasing L_(C), the velocity resolution improves at the expense ofrange resolution. Similarly, decreasing N and increasing L_(C) willimprove the range resolution at the expense of velocity resolution.

In some systems the signal has L_(C) chips per period but this sequenceis repeated M times and the correlation values are accumulated togenerate a signal complex sample for a given range. The sequence of suchsamples is then used for Doppler processing.

The above illustrates a tradeoff between the maximum unambiguous rangeand the maximum unambiguous velocity that only depends on the carrierfrequency. An increased product of unambiguous velocity and range canonly be obtained if the carrier frequency is decreased. In somecircumstances it might be desirable to obtain a larger unambiguous rangeat the expense of a smaller unambiguous velocity (or vice versa). Thus,a system that can adjust the repetition frequency of the signal would beable to adjust to different objectives. There is also a tradeoff betweenrange resolution and velocity resolution for a given bandwidth and scanduration. In some situations it would be advantageous to have betterrange resolution while in other cases it would be beneficial to havebetter velocity (or Doppler) resolution. Thus, it would be of benefit tobe able to adjust the system parameters depending on the objectivefunction of interest to obtain either the best range resolution or thebest velocity resolution (with a given fixed time interval for thescan).

As an example, consider a radar system with a desired scan duration(time to produce a velocity estimate) of 0.1 ms (100 scans per second).Suppose the chip rate is fixed at 10⁻⁸ seconds and the carrier frequencyis 79 GHz. A spreading code period of 100 chips would allow 1000repetitions in the scan time. This corresponds to an unambiguous rangeof 150 m and an unambiguous velocity estimate range of (−950 m/s, +950m/s). On the other hand, a spreading code period of 1,000 would allowonly 100 repetitions of the code in the same time. The unambiguous rangewould increase to 1,500 m, while the unambiguous velocity would decreaseto (−95 m/s, +95 m/s).

At the receiver it is necessary to store the complex outputs of thecorrelators for different possible ranges and for different receivers. Asequence of N complex samples needs to be stored for a particular rangeand a particular virtual receiver (a receiver matched to a particularspreading code of a transmitter) in order to determine an estimate ofthe velocity for an object at a particular range. For example supposethat there are 512 range bins desired to locate potential targets andthe number of repetitions of the code is 1024. This would requirestoring 512×1024 complex numbers with each complex number requiring 4bytes of storage. This would require more than 2 million bytes ofstorage per virtual receiver. If there are 4 transmitting antennas and16 receiving antennas then this would require about 134 Mbytes ofstorage, much more than is practical with current storage limitsintegrated onto a chip. On the other hand storing this off chip wouldrequire a significant amount of time to transfer data. At a rate of 1600Mbytes/second only about 12 transfers could happen per second. Thenumber of virtual receivers determines the possible angle resolution.More receivers can provide more angular resolution at the expense ofadditional storage or at the expense of worse range or velocityresolution. Thus, the storage restrictions limit either the angularresolution, the range resolution, or the velocity resolution.

In addition to the above, interference from other radar systems needs tobe accounted for. Interfering radars could be using the same type ofsignals as the vehicle in which the system of the present invention isinstalled. It is also possible that the interfering radar system isusing a different type of signal (e.g., FMCW vs. PMCW). It would beuseful to be able to mitigate in some way the effect of interferingradar systems. Different types of interference will require differentmitigation techniques. Mitigation of the effects of interfering systemsgenerally will not be ideal and it is often the case that themitigation, while reducing the effect of the interference, will alsodegrade the desired signal in some manner. If no interfering radarsystem is present, then it would be desirable to not employ themitigation technique. As such, it would be desirable to have a radarsystem that can adapt to the environment present.

In a preferred embodiment, the processing of the signals is shown inFIGS. 11 a, 11 b, and 11 c . FIG. 11 a illustrates exemplary processingmodules for a transmitter. A code generator 1102 generates a spreadingcode. The output of the code generator 1102 is modulated with a digitalmodulator 1104 to generate a complex baseband signal. The modulation isaccomplished in two parts. In the first part the code is mapped to acomplex sequence of in-phase and quadrature phase components at thedigital modulator 1104. The result is converted to an analog signal bythe digital-to analog converter (DAC) 1106. The output is further shapedwith a pulse shaper 1108 to generate a complex baseband analog signal.This signal is up-converted with a TX Mixer 1110. An oscillator 1124 isthe other input to the mixer to produce a radio frequency (RF) signal.The oscillator signal is also used at the receiver. This is indicated bythe connection of the oscillator to components in FIG. 11 b . The resultof up-conversion is then amplified by a power amplifier 1120 beforetransmission by an antenna 1122. A master clock 1126 is used to controlthe timing of the oscillator and to control the timing of the digitalcircuitry. The master clock 1126 and the oscillator are also shared withthe transmitter circuitry shown in FIGS. 11 b and 11 c . The output ofthe digital modulator 1104 is shared with the receiver so that thereceiver can apply interference cancellation. The output of the codegenerator 1102 is shared from the transmitter to receiver so appropriatecorrelation or matched filtering can be applied at the receiver.

FIG. 11 b illustrates exemplary analog processing circuitry of thereceiver. Various blocks or modules are illustrated. One or morereceiving antennas are connected to a switch 1142 that connects one ofthe antennas 1140 to a receiver. There can be more than one receiver sothat different antennas can be connected to different receivers. Not allthe antennas need to be connected to a receiver. Because there can bevery strong self-interference from the transmitted signal reflecting offof nearby objects (e.g., a bumper), the analog interference cancellationunit 1146 is employed. A signal from the cancellation unit 1146 can beprovided to the digital processing where additional interferencecancellation can be done. The output of the analog interferencecancellation 1146 is provided to a low noise amplifier 1148. The lownoise amplifier output is mixed down to baseband by an RF mixer thatalso uses the oscillator signal (from FIG. 11 a ). The resulting lowpass complex baseband analog signal is filtered (with low pass filter1152), and further amplified (with gain control 1154) before beingconverted to a digital signal by an analog-to-digital converter (ADC)1156. The result of the ADC 1156 is fed to digital processing circuitryshown in FIG. 11 c.

FIG. 11 c illustrates exemplary digital processing circuitry of thereceiver. Various signal processing blocks or modules are illustrated.First, a saturation detection block 1160 detects whether the ADC inputhas caused the ADC 1156 to saturate. This detection can be used toadjust the gain in the gain control 1154. Next, a change in the samplerate can be performed (1162) to reduce the amount of processingnecessary. After resampling, correction for any mismatch in I, Q gain ornon-orthogonality can be employed (via I/O Correction module 1164).Additional interference can be cancelled in a digital interferencecanceller 1166. Information from the processing done by the analogcancellation unit 1146 can be used (as shown by the connection from FIG.11 b ) by the digital interference cancellation unit 1166. This can moreaccurately (as compared to the analog interference canceller 1146)remove interference from near targets, including the bumper. Furtherinterference cancellation (with large target canceller 1168) can be doneto minimize the effect of sidelobes of a near target on thedetectability of a further target. Interference from other radarsystems, such as an FMCW system, can also be incorporated (such asFMCW/Tone Canceller 1170) into the digital processing. The resultinginformation is stored in a buffer 1174. This allows all digitalprocessing to be suspended temporarily in order to not create unwantedradio frequency interference from the digital processing. Finally, thesignal is processed by correlating with a correlator 1176, with delayedversions of the code from the code generator (1102). The correlator(s)1176 could be implemented in a number of ways including a matched filterand an FFT-based approach. The samples of the output of the correlatoror matched filter (1176) are stored in memory as radar data cubes (RDC),such as RDC1 (1178). The correlation values for different delays,different receivers, and different times are stored in the radar datacube. The information from RDC1 is processed further to determine objectvelocity and angle (e.g., azimuth or elevation or both). Furthersoftware control of the processing of information stored in RDC1 may beperformed to determine the velocity of targets.

The analog processing of the received signal from the antenna to the ADCis called the analog front end. The processing of digital signals fromthe ADC to RDC1 is called the digital front end. The processing ofdigital signals from the RDC1 to generate Doppler information and angleinformation is called the digital back end.

As mentioned above, the signals to be used for transmitting, and thereceiver processing to be employed, depend on a number of differentfactors including the environment (e.g., an urban area, suburban area,parking lot, garage, construction zone etc.). Different, changingobjectives for the radar system might be desired (e.g., small rangeresolution, small velocity resolution, small angular resolution, etc.).Different types of interference might be present in the radar system(e.g., FMCW radars, PMCW radars, etc.). Therefore, it is desirable to beable to dynamically adapt the radar to different environments, differentperformance objectives, and different types of interference. Embodimentsof the present invention provide for a software controllable adaptableradar system. An exemplary structure of the radar system is illustratedin FIG. 12 . The radar system will have a number of antennas 1202, 1230,transmitters 1220, and receivers 1200. In FIG. 12 , a number of antennas1202 are connected to a switch 1204. The switch 1204 allows the antennas1202 to be connected to a number of receivers 1200. In FIG. 12 , onlyone receiver 1200 is shown but there could be multiple receivers 1200. Areceiver 1200 will have an analog front end 1206, an analog-to-digitalconverter 1208, a digital front end 1210, and a memory 1212 for storingthe results of processing the signal that will be processed by a digitalback end. There could be a single analog front end 1206 and a single ADC1208 with multiple digital front end processing units 1210 andassociated memories 1212. In addition, a controller 1240 will be presentfor controlling the operation of the system. The controller 1240 willalso control the digital backend of the system. The controller willcomprise a control processor running software and memory storing thecontrol program. The memory used for the control processor could be partof a larger memory that also stores the information generated by thedigital front end 1210. The controller 1240 will control the digitalfront end 1210 and the analog front end 1226 of the transmitter 1220 andaspects of the analog-to-digital converter 1208.

The radar system will also include a number of transmitters 1220. Onesuch transmitter 1220 is shown in FIG. 12 . The transmitter 1220 willconsist of a digital front end 1222, a digital-to-analog converter (DAC)1224 and an analog front end 1226. It is also possible that antennas(1202, 1230) can be used for either transmission or reception (dependingon the configuration of the switch(es) (1204, 1228)).

FIGS. 13A and 13B illustrate the radar system architecture with multiplereceivers 1320. FIG. 13A illustrates one or more antennas 1300, followedby a switch 1310, followed by N_(T) receivers 1320, that is followed bymemory, such as radar data cube 1 (RDC1). The number of receivers 1320may be different than the number of antennas 1300. For example, a systemmight have 16 antennas 1300 but only 8 receivers 1320. In this case, 8of the antennas 1300 are not actually connected to a receiver 1320. Theswitch 1310 allows for any of the antennas 1300 to be connected to anyof the receivers 1320. The radar data cube (1330) stores outputs of eachreceiver 1320. The outputs are the correlations at a particular delay(range). One dimension of the radar data cube 1330 is the range ordelay, a second dimension is the virtual radar (transmitter and receivercode), and a third dimension corresponds to the sequence of complexcorrelator samples needed to calculate the velocity. For example, areceiver 1320 might correlate with one (or more) delays. The sequencecomplex correlation values will be stored in RDC1 (1330). Each of thereceivers 1320 in FIG. 13A will have an analog front end, ananalog-to-digital converter, and a digital front end. The digital frontend will, besides providing interference mitigation, performcorrelations with the spreading codes of different transmitters. FIG.13B illustrates an implementation of the correlator block (1176) of FIG.11 c . In FIG. 13B there are correlations performed with differentspreading codes corresponding to different transmitters. Thecorrelations can be done in many different fashions, such as with amatched filter that provides correlations with different delays. An FFTapproach can also be used whereby the input is transformed to thefrequency domain, as is the code. Then, multiplication followed by aninverse FFT operation is performed. Each of these methods producesoutputs for multiple delays. Finally, the correlation with a particulartransmitter (1340) can be accomplished with a multiply and sum operationwhere the product of the input signal and a particular delay of thetransmitted spreading code is generated and then summed over some windowof time. This would be repeated for various delays of the transmittedspreading code and would constitute one of the correlations with TX codeblocks (1340) in the receiver.

Self-Interference Mitigation

One aspect of this invention is self-interference cancellation.Self-interference refers to the effect of the signal from onetransmitter on the receiver/correlator matched to a second transmitterfrom the same radar system. U.S. patent application Ser. No. 15/481,648,now U.S. Pat. No. 9,689,967 (“the '648 patent application”, which isincorporated herein by reference in its entirety), describes a method ofself-interference mitigation. If there are multiple transmittersgenerating transmitted signals simultaneously, there will beinterference from transmitters to correlators matched to differenttransmitters. The method in the '648 patent application can be used tomitigate the interference at a receiver matched to one transmittedsignal due to signals transmitted by other transmitters (matched toother receivers). In the '648 patent application various modes ofoperation were considered. In one mode, only a single transmitter wasactively transmitting a signal. The receivers determined the effect ofthis transmitted signal on the output of the receivers matched to othertransmitted signals. In a second mode of operation, multipletransmitters are active. Depending on the environment a process ofselecting the modes of operation would be appropriate. This method ofself-interference mitigation can be controlled by a control processor todetermine which mode of operation should be employed and whichinterference should be mitigated at each receiver.

Other types of self-interference include spillover from a transmitter ona chip to a receiver on the same chip, antenna coupling, and reflectionsfrom fascia (e.g., bumpers) that can cause the analog front end of areceiver to operate in saturation. This “desenses” the receiver to thedesired signal from targets in the environment. The present inventionprovides a method for a phase modulated continuous wave (PMCW) systemusing the advanced temporal knowledge of the code(s) to be transmittedin combination with a bank of digital Finite Impulse Response (FIR)filters to generate complementary signal(s) to the self-interferencenoise, next convert them to an analog signal with a digital-to-analogconverter (DAC), and then subtract the complementary signals at one ormore points in the analog receive chain prior to desensing the receiver.This method can provide 20-40 dB of signal reduction. This significantlyreduces or eliminates the impact of these self-interference signals ondesensing the receiver Variable Gain Amplifiers (VGA).

In addition to desensing the receiver, these self-interfering signalsare still typically much larger in comparison to desired radar returnsignals even after the above signal reduction. These comparatively largeself-interference signals continue to cause issues in the digital domainfor detection of small targets at longer ranges. This interferencedegrades the signal-to-noise ratio (SNR) by raising side-lobes in range,Doppler, and angle.

The above analog cancellation method may also be used in the digitaldomain to recreate the digital version of the self-interferencesignal(s) with advanced temporal knowledge of the code(s) as inputs toan FIR filter bank and then digitally subtract the estimatedinterference signal in the digital domain prior to correlation andthereby lowering the impact of these by another 20-40 dB. Thisinterference mitigation technique can be combined with the multi-modalinterference mitigation technique described above.

Fascia Reflection Optimization

Integration of a radar system behind a plastic bumper of an automobile,or the fascia, can cause strong reflected signals from the transmitter.The fascia reflection and transmission loss varies greatly depending onthe layered composition of the fascia and the frequency of thetransmitted wave impinging on it. A radar system that has the ability toshift the carrier frequency of the transmitter and monitor the magnitudeof the received signal for the purpose of minimizing the effect offascia reflection can improve the performance of a radar system. Inaddition to minimizing the reflection, the frequency vs. reflectivitycan be determined and used in self-interference mitigation. Theselection of an appropriate carrier frequency is controlled by a searchalgorithm implemented in hardware or software running on a controlprocessor. Using this, the frequency that provides the maximumsensitivity can be found.

Quadrature Orthonormal Calibration

Another aspect of a radar system that can be controlled by a processoris balancing the in-phase and quadrature mixing process at the receiver.Quadrature receivers rely on perfectly orthonormal signal processingpaths, one for the I channel, and the other for the Q channel. Practicalimplementation of the two paths suffers from both magnitude (gain) andphase inaccuracies. The gain or amplitude mismatch between the I and Qchannels is compensated using a method well practiced in the art. Thephase mismatch between the I and Q channels, which corresponds to arelative I vs. Q phase different from the desired 90 degrees, is moredifficult to compensate and usually requires a significant amount ofhardware. The raw input data to the correlators or matched filters needsto be adjusted in regards to gain and amplitude. This is done insubsystem (1164) shown in FIG. 11 as part of the receiver pipeline. Inaddition, the phase has to be corrected and is also performed insubsystem (1164). Since the analog section performs some automatic gaincontrol, the output of the correlators needs to be adjusted accordingly.The gain control can move the individual correlation values (I or Q) upor down by 1 bit. The gain at any time can be read out from the analogsection by the control processor, and that gain number can be appliedfor the full duration of a correlation (Lc chips). The gain control hasto be performed for each physical receiver (for I and Q separately).Based on the binary representation of the gain control, the values areshifted by one to the left or right, or no changes are required.

The multiplication factor is provided by the control processor and willresult in a hardware multiply (16 bit). This has to be done for both Iand Q. The I/O correction ensures that the I/O values are fullyorthogonal, and therefore adjusts the angle and amplitude.

The required adjustments can change based on temperature but also basedon the iterative decoding adjustments, and therefore these adjustmentsneed to be computed completely (which requires trigonometric functions)and cannot be pre-computed by the control processor. Phase correction isrequired to compensate for the different wire lengths and antennamismatch.

The present invention compensates for both the I/O gain and phasemismatch. This is accomplished by a control processor utilizing thecorrelation values, input from both the analog processing system and thecorrelator outputs in order to provide orthogonal and equal gain I and Qchannels. Both of these compensation methods can be controlled by theprocessor running software stored in memory.

Resolving Range and Doppler Aliasing

The present invention also provides a PMCW radar using periodicsequences for pulse compression, which requires many interdependentparameters to be set correctly in order to achieve a particular set ofperformance requirements. These parameters include the following.

i. T_(c)—Chip time

ii. L_(c)—Sequence length, in chips

iii. M—Number of repetitions of sequence in coherent accumulation togenerate a single point for Doppler processing

iv. N—Number of Doppler sampling points.

The product of these 4 numbers is known as T_(d), the dwell (or scan)time. The maximum unambiguous distance that can be resolved is D_(u)=(cT_(c) L_(c)/2). Due to other constraints in the system, which imposelimits on T_(d), M, and N, it is often impractical or impossible to setT_(C) and L_(C) large enough to achieve the desired D_(u). In suchcases, targets at a distance D greater than D_(u) will alias into arange between 0 and D_(u) that is equal to D modulo D_(u).

Similar aliasing also happens in Doppler velocity as discussed above.The present invention provides a method for resolving range and Doppleraliasing, comprising:

-   -   a. Performing 2 or more consecutive scans with different D_(u),        e.g.: using a different L_(C) or T_(C);    -   b. Detecting targets in said scans, and determining aliased        range and Doppler velocity for said targets; and    -   c. For each detected target in the scans, calculate its expected        aliased range (and Doppler) for 0 . . . N levels of folding in        every other scan, and select the level of folding for which        there is a corresponding detection in all scans.

The above steps are controlled by a processor running software whichallows for various unambiguous ranges and unambiguous velocities.

Removing FMCW Interference on Radar

Current FMCW Radars have low capabilities to cancel out other FMCWRadars. Current methods include:

-   -   a. Enabling random frequency sweeps and detection and dropping        of scans.    -   b. Fixing frequency overlap points by dropping and estimating        value.    -   c. Typical systems can handle a handful of other interfering        radars.

The present invention provides a method of broadband acquisition thatenables detection and digital subtraction of the interfering FMCWsignals. The implementation uses multiplicity of an adaptive tonetracking system to lock on the largest tones and then sequentialdigitally cancel them prior to bandwidth compression. The acquisition ofinterference takes on the order of 10 to 20 samples and are outside ofthe bandwidth of interest. Once a tracking error is low enough, theinterfering signal can be directly subtracted out of the time datastream. After the interfering signals are removed, the signal isfiltered and compressed to the desired band. Note, the bandwidthcompression naturally enables bit growth of the sampled signal.

The present invention's interference cancelation can be used with eitherPMCW or FMCW systems. For example, an ADC can sample at 3.5 GHz with 6.2effective number of bits (ENOB). Typical FMCW systems sample at 10 MHzwith 10 ENOB ADC. In accordance with the present invention, theinterferers can be subtracted out and then compressed by a factor of3500/10 or 350× with a real bit growth of 4.2 or total of 10.4 bitswithout any interference. This exemplary system can track and remove anynumber (>100) of interferers with little impact to SNR or distortion toexisting signals.

PMCW Frequency Scanner

The present invention provides for scanning the available bandwidth anddetermining a noise floor at each frequency, thereby making it possibleto identify low interference sections of the band and to adjust thecenter frequency and the bandwidth of the radar scan to minimize theimpact of the interferers. The present invention includes a processorrunning software controlling the adjustment of the frequency and thebandwidth of the transmitted signals in a radar system.

Reducing Interference from PMCW to PMCW Radar

PMCW radar relies on the use of PRN sequences to be undetectable tothose who do not have the ability to match a received filter to thetransmitted waveform. However, there is nevertheless the possibility ofinterference because of mismatched interference power to desiredreflection power. Reducing interference is important since it isexpected that the percentage of cars using radar will increasesubstantially. Mitigating PMCW interference in PMCW radar systems isdiscussed in U.S. patent Ser. No. 15/416,219, filed on Jan. 26, 2017,now U.S. Pat. No. 9,772,397, where is hereby incorporated by referenceherein in its entirety.

To reduce the interference from other automobiles using the PMCW radarof the present invention, the system employs a number ofcounter-measures. The following is a list of exemplary measurements andcounter-measures that the system may employ:

-   -   a. Use of outer codes that are orthogonal to transmitted codes        for a continuous measurement of interference. Once interference        has been determined to be too large, countermeasures for dealing        with interferers may be implemented.    -   b. Offsetting the center frequencies by a certain amount that is        larger than the expected signal from the fastest Doppler. For        example, if a Doppler of 40 KHz could be expected and a Doppler        sampling rate of 120 KHz is selected, separating the center        frequencies by 6× the sampling rate will sufficiently        decorrelate other interferers. This is accomplished by changing        the center frequency of the transmitter and receiver along with        randomizing the clocks. For the system of the present invention,        the system may have a 10 MHz center frequency selection and may        enable 100 PPM crystals for the reference clock. The system can        spread the center frequencies between 76 to 81 GHz in 10 MHz        increments with a +/−7.9 MHz sub distribution due to the        crystals.    -   c. Scan the entire frequency spectrum and determine where the        noise is the lowest and place the next scan in that section.    -   d. Use many virtual receivers to isolate the interferer into a        small angle. The system may increase the number of time        interleaved VR's to the maximum to increase isolation.    -   e. Switch to codes that are more robust to interference such as        a pseudorandom binary sequence (PRBS) with randomization. The        interfering radars will show up in the noise floor.    -   f. Use several smaller scans of PRBS or non-identical codes,        process each through beamforming and determine which targets are        ghosts and which ones are persistent. The ghosts are eliminated        and the persistent ones kept.

The selection of which of these techniques to use is controlled by aprocessor running software, such that a particular interferencemitigation technique may be selected based on the current operationaland environmental conditions.

Hadamard Noise Floor Inspection

The present invention also provides for generating more Hadamard codesthan used in a conventional multiple-input, multiple-output (MIMO) radarsystem. The unused codes may be correlated to the received signal todetermine the interference level of the current scan. This is possiblebecause all of the transmitted codes should be orthogonal to the extracode. Any rise in the noise floor will be from other radars transmittingat different center frequencies and/or different codes. This can be usedto determine if a scan abort or other counter measures will be needed tobe deployed.

Adaptive Transmission and Interference Cancellation for MIMO Radar

As discussed in detail in U.S. patent application Ser. No. 15/481,648,filed Apr. 7, 2017, now U.S. Pat. No. 9,689,967, which is herebyincorporated by reference herein in its entirety, an exemplary MIMOradar system has different modes of operation. In one mode the radaroperates as a SIMO system utilizing one antenna at a time. Codes withexcellent autocorrelation properties are utilized in this mode. Inanother mode the radar operates as a MIMO system utilizing all theantennas at a time. Codes with excellent cross correlation propertiesare utilized in this mode. Interference cancellation of the non-idealautocorrelation side lobes when transmitting in the MIMO mode areemployed to remove ghost targets due to unwanted side lobes.

There are several types of signals used in radar systems. One type ofradar signal is known as a frequency modulated continuous waveform(FMCW). In this system the transmitter of the radar system transmits acontinuous signal in which the frequency of the signal varies. This issometimes called a chirp radar system. At the receiver a matched filtercan be used to process the received signal. The output of the matchedfilter is a so-called “pulse compressed” signal with a pulse durationinversely proportional to the bandwidth used in the chirp signal.

Another radar signal is known as a phase modulated continuous waveform(PMCW). In this system the phase of the transmitted signal is changedaccording to a certain pattern or code also known at the radar receiver.The faster the phase is changed, the wider the bandwidth of thetransmitted signal. This is sometimes called spread-spectrum because thesignal power is spread over a wide bandwidth. At the receiver a matchedfilter is used that produces a so-called pulse compressed signal as wellwith time resolution proportional to the inverse bandwidth of thetransmitted signal. Codes with good autocorrelation values are importantin phase modulated continuous wave radars.

Radars with a single transmitter and single receiver can determinedistance to a target but cannot determine the direction of a target. Toachieve angular information either multiple transmitters or multiplereceivers or both are needed. The larger number of transmitters andreceivers, the better resolution possible. A system with multipletransmitters and multiple receivers is also called a multiple-input,multiple-output or MIMO system. One method of canceling outself-interference is to generate a replica of each spreading code at thereceiver. This signal is then used as an input to an FIR filter thatwill reconstruct the received signal corresponding to the transmittedsignal of user i. By just inverting this signal and adding it to theinput of the filter matched to the j-th transmitted signal, the i-thtransmitted signal will be automatically removed. By updating the tapsof the FIR filter as the vehicle moves, the interference will besignificantly reduced.

Implementing the FIR filter can also be done in the frequency domain bytaking the FFT of the replica of the spreading code of user i,processing it (multiplying) with the FFT of the spreading code of user jand then further multiplying it by the known channel characteristics. Assuch, the part of the received signal due to user i can be recreated atthe receiver attempting to process user j's signal. Once recreated, thissignal can be used to cancel out the signal of user i. Note that thegeneration of the correlation between the signal of user i and that ofuser j can be used at all the receivers but only needs to be generatedonce. The benefit of this approach is that codes that have goodautocorrelation but potential poor cross correlation will not cause aproblem with the system. As such a search for codes with goodautocorrelation (such as m-sequences, APAS sequences) would besufficient.

The present invention provides a method of using MIMO radar in which thetransmitted signal adapts based on the current knowledge of targets. Atturn on, with no knowledge of the targets, the radar will use oneantenna at a time (SIMO mode). A sequence with excellent autocorrelationproperties (e.g., m-sequences, APAS sequences, Golay sequences, and thelike) is employed initially by a single transmitter. The recoveredsignal is processed to determine a coarse range estimate and possibly aDoppler estimate for each target. This might involve a combination ofcoherent integration and noncoherent integration depending on the rangeof Dopplers anticipated. Each of the individual antennas is sequentiallyused.

After each of the transmitters has been used once and coarse knowledgeof range is available, then the system switches to MIMO mode in whichall transmitters are used simultaneously. In this mode sequences withgood cross correlation are utilized. The nonideal properties of theautocorrelation of these sequences, can be neutralized by interferencecancellation techniques. One embodiment uses m-sequences for the SIMOmode and uses a combination of m-sequences and Hadamard codes for theMIMO mode. Another embodiment uses APAS codes for the initial sequencesand a combination of Hadamard codes and APAS codes for the MIMO mode. Athird embodiment uses Golay codes (with QPSK) for the SIMO mode andHadamard codes for the MIMO mode. Different interference cancellationtechniques can be employed for the MIMO mode to eliminate (or reduce)the interference from side lobes of the autocorrelation of thesequences.

These techniques are controlled by a processor running software thatallows the radar system to dynamically adapt to the environment, thedesired performance criteria, and external inputs.

Improving Processing Gain by Shifting the Doppler Estimation

Radar velocity estimation resolution can be improved, as described inU.S. provisional application, Ser. No. 62/327,016, which is herebyincorporated by reference herein in its entirety. An automotive radarrequires the ability to discriminate targets moving at a relativevelocity of −250 kph to +500 kph. The hardware detector that processesthe Doppler frequency shift created by the relative velocity of thetarget is usually built as a symmetrical system, designed to handlerelative velocities from −500 kph to +500 kph. As a consequence, thesignal processing calculations treating the range from −500 kph to −250kph is wasted (since it is not necessary to process data on an objectmoving 250 kph or more away from the subject vehicle). The presentinvention pre-processes the signal entering the Doppler estimator suchthat the same symmetrical design can treat velocities from −250 kph to+500 kph with no wasted computation cycles on unused velocity ranges.

The preprocessing comprises the steps of: (i) determining, for eachsample of the received time-series of complex (In-Phase & Quadrature)samples, the phase shift necessary to produce a required Dopplerfrequency shift in the output of the Doppler Processing (e.g., FFT orChannelizer), and (ii) multiplying each complex sample of the capturedtime-series by a matrix to rotate the phase angle by that amount. Asignificant advantage of this technique is the fact that it effectivelyimproves the Doppler velocity resolution for the same number ofaccumulated sensed points. Furthermore, if the preprocessing of the datasets is done before the pulse compression engine, compression levels canbe increased further for corresponding increased processing gain.

Use of Time-Multiplexed Radar Scans to Reduce HW Overhead and to EnhanceDetection Quality

The present invention provides for the use of fully re-configurableradar scans (antenna pattern, frequency, LC, M, N, FFT points, range bininterval and the like) which can be used to focus on different area ofinterests. This allows the radar to adapt to different scenarios (e.g.,parking, vs driving). Different radar scans can also be configured toextend the range by ensuring that one radar scan, for example, one radarscan scans from 0-30 m while another radar scan scans from 30-200 m.Radar scans can also be used to focus on points of interests within thefield of view. Different radar scans could also be used to disambiguatebetween targets, which might have aliased back. The use of differentradar scans allows the system to keep the HW (hardware) requirements(e.g., memory) fairly small while being able to adapt to differentscenarios or even to focus on certain areas of interest. For differentframes, or even within a frame, different radar scans with short andlong dwell times can be used.

Another example is that if one radar scan used all 32 virtual receives(VR) with 80 range bins for high angular accuracy and the next one (ormore) used just 8 VR with 320 range bins for more distance resolution.This can be combined with another scan for better velocity resolution.

The present invention thus provides flexibility in adjusting radar scansto focus on different areas of interest and to be able to use severalrange scans to virtually increase the resolution. The adjustment ofradar scans is controlled by the processor running software that candynamically adapt to the environment, the desired performance criteria,and external information.

Radar Data Compression

In a radar system, Doppler processing typically comprises performing aFast Fourier Transform (FFT), a sufficiently long time series of complexdata. This Doppler processing is performed independently on multipletime series, captured simultaneously by multiple virtual receivers formultiple range bins. However, the entire time series must be availablebefore processing can begin. This requires partial time series to bestored in memory as the points of the time series are captured, whichcan result in a large amount of memory being used. In order to enablelonger scans, or scans with more virtual receivers or more range bins,using a limited amount of memory, it is desirable to first compress thetime series as they are captured, then store the compressedrepresentations in memory, then once they are fully captured, decompressthem prior to performing Doppler processing on them. The compressionshould be lossless to avoid introducing compression artifacts into thedata.

The present invention proposes a method and device for losslesslycompressing radar data cube (‘RDC1’) data prior to performing Dopplerprocessing, by using one or more predictors to reduce the number of bitsthat are required to represent each complex number of the time series.Data may be optionally converted from complex in-phase and quadrature(I/Q) format to phase angle and magnitude format prior to the predictionstep.

After prediction, residual values (the difference between the predictedvalue and the actual value) are compressed using an arithmetic orentropy encoding algorithm. Within a given range bin, predictors predictsamples of data based on the value of samples from adjacent virtualreceivers (to either side of the sample being predicted), and fromprevious time samples for the same virtual receiver. Inter-range binprediction may also be used.

Compensating for Doppler Shift

Doppler shift has an adverse effect on a radar system's ability tocorrelate a received signal with various shifts of the transmittedsignal to determine range. As the phase of the received signal rotatesdue to Doppler shift, the magnitude of the zero-shift peak decreases,and the average magnitude of the non-zero-shift “sidelobes” increases.While the former lowers the SNR for the target in question, the latterlowers the SNR for all targets in all other range bins (non-zeroshifts). Naturally, largest target(s) cause the largest sidelobes.

The present invention provides a method and device for compensating forDoppler Shift, comprising:

-   -   a. Identifying a set of the largest (highest received signal)        targets;    -   b. Calculating the central the Doppler velocity for the largest        targets:    -   i. Median, weighted arithmetic mean, weighted geometric mean,        and the like;    -   c. Calculating the phase shift, X, corresponding to said median        Doppler velocity; and    -   d. Prior to correlation, rotating the phase of the Nth sample of        the coherent integration by N*X.

The present invention is a control processing unit that dynamicallycontrols the signal processing described above depending on theenvironment, the performance criteria, and external information.

An Increased Entropy PRNG

The present invention provides a method to increase the randomness of analgorithmically generated pseudorandom binary (PRB) sequence, forexample, LFSR based, where true thermal noise, already present at thequantizer output of the radar receiver, is used. The thermal noise inthe radar receiver path is much larger than the size of the LSB of thequantizer present at the backend of the receiver path. As a consequence,the quantizer LSB is switching between 1 and 0 in a truly randomfashion. Exemplary implementations use the random nature of thequantizer LSB and imprint it on the algorithmically generated PRBsequence by means of an XOR operation. The entropy can be furtherimproved by using the same XOR operation against the LSBs of allavailable quantizers in the system. This enables the generating of acode that cannot be predicted and decreases the severity of maliciousjammers to only replay attacks that can generate false targets furtherthan the jammer. This also limits the extent and scope of potentialfalse operations due to the jammer.

The random number generator is dynamically controlled by a processorthat depends on one or more of the environment, the performancecriteria, and external information. Increased Entropy PRNG is alsodiscussed in U.S. Pat. No. 9,575,160, and U.S. provisional applicationNo. 62/327,017, which are hereby incorporated by reference herein intheir entireties.

Range Walking/Sub Range Resolution in Analog or Digital Domain on theTransmit and Receive Side

In PMCW radar, the chip frequency (i.e., the modulation frequency)determines the range (radial distance) resolution. For example, a 2 GHzchip frequency normally achieves a range resolution of around 7.5 cm.The present invention provides a method and device for Range Walking,and enables a higher range resolution without having to use a higherchip frequency. In order to achieve finer resolutions without increasingthe frequency, several consecutive scans are performed, each with adifferent sub-cycle transmit delay in the code (for example, multiplesof 1/32 of a clock cycle).

On the receive side, correlation is performed with the original (i.e.:not delayed) code sequence. This progressive delay across scans allowsclosely spaced targets (i.e.: less than one range bin) to be resolvedafter multiple scans. For example, a multiple of 1/32 cycle delay with a2 GHz chip frequency would achieve about 0.25 cm resolution. Rangewalking is also possible on the receive side with an FIR resamplingfilter, comprising the steps of operating the Analog to DigitalConverter (ADC) at a higher sampling rate than the chip frequency, andusing different resampling filters to shift in time the center point ofthe filter by sub-chip intervals.

Furthermore, combining transmit-side range-walk and receive-sideresampling with different relatively prime sub-chip time shifts can beused to give a greater number of subchip delays: for example, range-walkdelays of m/M chip (where m is an integer from 0 to M−1) and resamplingdelays of n/N chip (where n is an integer from 0 to N), where N and Mare relatively prime, results in a total of MN different shifts beingavailable to achieve maximum range resolution.

Multi-Chip Radar

In PMCW systems, increasing the number of virtual receivers in each chipcan be accomplished by correlating to the transmitters from the otherchip(s). This is enabled through the distribution of several signals:start of scan, phase information distributed by means of a clock, andthe codes transmitted. For PRBS codes, seeds and taps that are beingused can be shared to synchronize the chips, for example, for the codestransmitted. If these chips are in close proximity (in a same housing),they can be assumed to be a single system and increases the VRfunctionality. When the chips are located in a separate housing, longbase line interferometry and triangulation is used for additionalangular resolution. For example, the chip of the system of the presentinvention may include, for example, 12 transmitters (Tx) and 8 receivers(Rx), and the system can use 12 Tx from the other chip, as long as thetransmitters are phase aligned, and the system knows the PRN sequencewhich is being transmitted from the other chip. For the one chip thiswould look like the system is using 24 Tx and 8 Rx (192 VirtualReceivers)—the same applies for the other chip—so instead of 2×96 VRs,the system actually gets 2×192 VRs with two chips. This can of coursescale assuming the different chips have the HW to deal with theadditional virtual receives.

Antenna Switches

The present invention also provides for the use of antenna switches toincrease the number of Virtual Receivers/angular resolution as well asto adapt to different antenna patterns (e.g., for LRR vs parking radar).The use of antenna switches on either the transmitter or receiver allowsa single radar chip to time-multiplex different antenna patterns andperform radar scans, e.g., for long range radar and short range radarwith different antenna characteristics. The use of antenna switches alsoallows an interleaved antenna mode, which can be used for additionalspatial resolution.

The control of the antenna switches is accomplished by a controlprocessing unit that dynamically adapts to factors including theenvironment, the desired performance criteria, and external information.

Method to Pulse PMCW Radar and Power Shape

In a PMCW radar, auto-correlation of code sequences is used to measurerange to targets. When using codes that do not have perfect (zerovalued) off-peak auto-correlation, a vehicle's bumper (through which anautomotive radar must typically operate) as well as large nearbytargets, can cause large side lobes in all the range bins. These sidelobes are detrimental to good radar performance as they can easily hidesmaller targets, especially at greater ranges.

The present invention proposes a method and device for eliminating orreducing the adverse effect of very close reflectors by alternatingbetween transmit-only operation and receive-only operation, optionallywith a short delay (Q) between transmitting and receiving. In oneembodiment, a maximum range of interest is selected, and one round triptime of the signal (at the speed of light) from the radar to the targetand back to the radar is used as the pulse length (T). For a period oftime T-Q, the transmitter transmits a code sequence (with the receiverturned off). Then the transmitter is turned off for time Q. Next, thereceiver is turned on for time T. Then the pattern is repeated. Theeffect of this scheme is that the receiver does not receive any signalfrom nearby targets for which the round-trip time is less than Q.Furthermore, the receiver receives the maximum signal from targets atthe maximum distance (i.e., for time T-Q). For a target at anintermediate distance, the time during which the receiver receives asignal from said target is proportional to the distance (i.e., thegreater the distance, the more time the signal is received).

In another embodiment, the transmit power is continuously adjustedduring the transmit period (T−Q) as follows: at the beginning of theperiod, maximum transmit power is used, whereas at the end of theperiod, zero (or minimal) transmit power is used. Various power curves(power vs. time) can be implemented. For example ((T−t)/T)³, where “t”goes from 0 to T, will negate the effect of range in the Radar Equation(i.e., K/R⁴), resulting in equal received power for equal sized targetsat different ranges (up to the maximum range of interest). Such pulsedand power shaped RF signals are discussed in detail in U.S. patentapplication Ser. No. 15/292,755, filed Oct. 13, 2016, now U.S. Pat. No.9,753,121, which is hereby incorporated by reference herein in itsentirety.

FIR Down-Sampling

In the receive side processing for a PMCW radar, it is desirable to beable to change the chip (phase modulation) frequency without having tochange the actual analog clock frequency, which typically requiressignificant time to change. The present invention proposes a method anddevice for down-sampling from a fixed sampling frequency to a lowermodulation frequency, for example, to 500 Mhz from 2 GHz, comprising thesteps of running the Analog to Digital Converter (ADC) at a fixedfrequency, greater-than-or-equal to the modulation frequency, andfiltering samples from the ADC to produce samples at a lower modulationfrequency. Optionally, an FIR filter is used to perform the filtering.

Quiet/Delay Buffer

The switching of the digital components on the chip can cause additionalnoise levels, which preferably is removed especially if looking forobjects relatively far away. In order to do that the system of thepresent invention can limit the digital components by storing theincoming data from the analog section in a buffer and processing thatdata a little bit delayed after all the important information isreceived. This ensures that the system can limit the digital noise toimprove the signal-to-noise ratio especially when the system is tryingto detect far objects.

The quiet buffer can also be used as a delay buffer, e.g., the incomingsignal can be stored at 2 GHz while the processing can be performed at 1GHz.

A Quiet buffer/delay buffer can be used to quiet down the chip whilereceiving the return from far objects as well as used to reducefrequency requirements on the digital components.

Power Shaping of Single Antenna

The present invention provides for the use of a multiple strip or slotantenna arrays, which makes it possible to create an antenna power shapethat will cover both long, medium and short range targets. This isaccomplished through constructive and destructive interference of two ormore candelabra arrays. This reduces hardware requirements by 2-3× overdeployed embodiments.

Changes and modifications in the specifically described embodiments canbe carried out without departing from the principles of the invention,which is intended to be limited only by the scope of the appendedclaims, as interpreted according to the principles of patent lawincluding the doctrine of equivalents.

The invention claimed is:
 1. A method for operating a radar sensingsystem, the method comprising: configuring a transmitter to transmit aradio signal; configuring a receiver to receive radio signals, whereinthe received radio signals include the transmitted radio signaltransmitted by the transmitter and reflected from objects in theenvironment; with advanced temporal knowledge of the codes used tomodulate the transmitted radio signal, using code values of theplurality of codes, in combination with a bank of digital finite impulseresponse (FIR) filters, generating complementary signals of anyself-interference noise, and subtracting the complementary signals atone or more points in the receiver prior to the interference desensingthe receiver.
 2. The method of claim 1, wherein the receiver comprises abuffer, and further comprising storing received analog signal data intothe buffer, and processing, with the receiver, the received analogsignal data stored in the buffer after the analog signal data has beenreceived.
 3. The method of claim 1 further comprising delaying digitalprocessing the received analog signal data to avoid creating unwantedradio frequency interference from digital processing.
 4. The method ofclaim 1, wherein the transmitter comprises a digital processing unit, adigital-to-analog converter, and an analog processing unit, and whereinthe receiver comprises an analog processing unit, an analog-to-digitalconverter, and a digital processing unit.
 5. The method of claim 4,wherein the digital processing unit changes processing depending on atleast one of changing environment, changing objectives, and changingtypes of interference.
 6. The method of claim 5, wherein the changingenvironment comprises an operational area, and wherein the objectivescomprise one of range, velocity, and angular resolution.
 7. The methodof claim 1, wherein the transmitter modulates the transmitted radiosignal as defined by a plurality of codes.
 8. The method of claim 7,wherein the transmitter transmits modulated continuous-wave radiosignals.
 9. The method of claim 1, wherein the radar sensing systemcomprises a plurality of receivers and a plurality of transmitters eachconfigured for placement in a vehicle.
 10. A radar sensing systemcomprising: a transmitter configured to transmit a radio signal; and areceiver configured to receive radio signals, wherein the received radiosignals include the transmitted radio signal transmitted by thetransmitter and reflected from objects in the environment, and whereinthe receiver comprises a frequency modulated continuous wave (FMCW)interference canceller configured to detect the largest interferencesignals and sequentially cancel them while signal processing thereceived radio signals.
 11. The radar sensing system of claim 10 furthercomprising a buffer, wherein the receiver is configured to storereceived analog signal data into the buffer, wherein the receiver isconfigured to process the received analog signal data stored in thebuffer after all the analog signal data has been received.
 12. The radarsensing system of claim 11, wherein the receiver is configured to delaydigital processing of the received analog signal data to avoid creatingunwanted radio frequency interference from digital processing.
 13. Theradar sensing system of claim 10, wherein the transmitter comprises adigital processing unit, a digital-to-analog converter, and an analogprocessing unit, and wherein the receiver comprises an analog processingunit, an analog-to-digital converter, and a digital processing unit. 14.The radar sensing system of claim 13, wherein the receiver is operableto change the processing of the digital processing unit depending on atleast one of changing environment, changing objectives, and changingtypes of interference.
 15. The radar sensing system of claim 14, whereinthe changing environment comprises an operational area, and wherein theobjectives comprise one of range, velocity, and angular resolution. 16.The radar sensing system of claim 10, wherein the transmitter isconfigured to modulate the transmitted radio signal as defined by aplurality of codes.
 17. The radar sensing system of claim 16, whereinthe transmitter is configured to transmit modulated continuous-waveradio signals.
 18. The radar sensing system of claim 10, wherein theradar sensing system comprises a plurality of receivers and a pluralityof transmitters each configured for placement in a vehicle.
 19. Theradar sensing system of claim 10, wherein the receiver comprises adigital interference canceller and an analog interference canceller,each configured to remove self-interference.